Clocked inverter, nand, nor and shift register

ABSTRACT

A threshold voltage of a transistor is fluctuated because of fluctuation in film thickness of a gate insulating film or in gate length and gate width caused by differences of used substrates or manufacturing steps. In order to solve the problem, according to the present invention, there is provided a clocked inverter including a first transistor and a second transistor connected in series, and a compensation circuit including a third transistor and a fourth transistor connected in series. In the clocked inverter, gates of the third transistor and the fourth transistor are connected to each other, drains of the third transistor and the fourth transistor are each connected to a gate of the first transistor, sources of the first transistor and the fourth transistor are each electrically connected to a first power source, a source of the second transistor is electrically connected to a second power source, and an amplitude of a signal inputted to a source of the third transistor is smaller than a potential difference between the first power source and the second power source.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a clocked inverter and also relates toa shift resister including a clocked inverter as a unit circuit.Further, the present invention relates to electric circuits such as aNAND and a NOR.

2. Description of the Related Arts

In recent years, display devices such as a liquid crystal display deviceand a light emitting device have been developing greatly because of thegrowth in demand of mobile machines. A technique for integrating a pixeland a driver circuit (hereinafter, internal circuit) using a transistorformed of a polysilicon semiconductor on an insulator has beendeveloping greatly, because the technique can contribute tominiaturization of devices and less electric power consumption. Theinternal circuit formed on an insulator is connected with a controllerIC or the like (hereinafter, external circuit) thorough a FPC or thelike to be controlled.

Generally, the power source voltage of an internal circuit isapproximately 10 V whereas an IC that constitutes an external circuitprepares a signal with approximately 3 V amplitude, since the IC canoperate with lower power source voltage than an internal circuit. Inorder to accurately operate an internal circuit with the signal withapproximately 3 V amplitude, there is a shift register in which a levelshift portion is arranged in each stage. (Reference 1. Japanese PatentLaid-Open No. 2000-339985)

FIGS. 11A, 11B, 11C and 11D show a circuit diagram of a clockedinverter, a logic symbol of the clocked inverter, a circuit diagram of aNAND and a circuit diagram of a NOR, respectively.

When level shifting is performed in an internal circuit, problems arecaused, for example, in increase in occupation area of a driver circuit,reduction of frequency property due to delayed or blunted waveforms.Furthermore, as described in the Reference 1, it is necessary tosuppress fluctuation in TFT characteristics between adjacent TFTs whenthe current driving type of shift register is used. On the contrary,when a level shifter is arranged in an external circuit, problems arecaused, for example, growth in total size of a casing for devices due tothe increase in the number of components such as IC, in cost formanufacturing and in power consumption by the shift register.Accordingly, it is preferable to use a signal with approximately 3 Vamplitude without level shifting.

Further, a threshold voltage of a TFT is fluctuated because offluctuation in film thickness of a gate insulating film or in gatelength and gate width caused by differences of used substrates ormanufacturing steps, and thus the threshold voltage value may bedifferent from an expected value. In such case, when a signal with asmall amplitude, approximately 3 V amplitude is used in a digitalcircuit in which two logical level, 1 and 0 are used, the TFT may not beoperated accurately due to the influence of the fluctuation in thethreshold voltage.

SUMMARY OF THE INVENTION

The present invention has been made in view of the above problems. It isan object of the present invention to realize miniaturization of acasing for devices and to reduce manufacturing costs and powerconsumption by providing the shift register without arranging any levelshifter in an external circuit. Further, according the presentinvention, the shift register can be achieved without arranging anylevel shifter in an internal circuit to solve such problems that thewaveform of CK is delayed and blunted and that the voltage of a powersource line arranged in the internal circuit is dropped. Also, thereduction of an area occupied by a driver circuit in the internalcircuit, the reduction of power consumption, and a high frequencyoperation can be realized.

Further, it is another object of the present invention to provide aclocked inverter, a shift register that can be operated accurately bymitigation of the influence of the fluctuation in the property of TFT.Moreover, it is possible to provide a NAND circuit or a NOR circuit thathas lower input load and higher output ability as compared toconventional NAND circuit or NOR circuit.

In order to achieve the above-mentioned objects, according to thepresent invention, there are employed the following measures.

According to the present invention, there is provided a clocked inverterincluding:

a first transistor and a second transistor connected in series, and

a compensation circuit including a third transistor and a fourthtransistor connected in series, in which:

gates of the third transistor and the fourth transistor are connected toeach other;

drains of the third transistor and the fourth transistor are eachconnected to a gate of the first transistor;

sources of the first transistor and the fourth transistor are eachelectrically connected to a first power source;

a source of the second transistor is electrically connected to a secondpower source; and

an amplitude of a signal inputted to a source of the third transistor issmaller than a potential difference between the first power source andthe second power source.

According to the clocked inverter of the present invention, the firstpower source is a high potential power source, the second power sourceis a low potential power source, the first transistor and the fourthtransistor are each a P-type transistor, and the second transistor andthe third transistor are each an N-type transistor.

According to the clocked inverter of the present invention, the firstpower source is a low potential power source, the second power source isa high potential power source, the first transistor and the fourthtransistor are each an N-type transistor, and the second transistor andthe third transistor are each a P-type transistor.

According to the present invention, there is provided a NAND including:

a first transistor and a second transistor connected in parallel;

a third transistor connected to the first transistor and the secondtransistor in series; and

a compensation circuit including a fourth transistor and a fifthtransistor connected in series, in which:

gates of the fourth transistor and the fifth transistor are connected toeach other;

drains of the fourth transistor and the fifth transistor are eachconnected to a gate of the third transistor;

sources of the first transistor and the second transistor are eachelectrically connected to a high potential power source;

sources of the third transistor and the fifth transistor are eachelectrically connected to a low potential power source; and

an amplitude of a signal inputted to a source of the fourth transistorand each of gates of the first transistor, the second transistor, thefourth transistor, and the fifth transistor is smaller than a potentialdifference between the high potential power source and the low potentialpower source.

According to the present invention, there is provided a NOR including:

a first transistor and a second transistor connected in parallel;

a third transistor connected to the first transistor and the secondtransistor in series; and

a compensation circuit including a fourth transistor and a fifthtransistor connected in series, in which:

gates of the fourth transistor and the fifth transistor are connected toeach other;

drains of the fourth transistor and the fifth transistor are eachconnected to a gate of the third transistor;

sources of the first transistor and the second transistor are eachelectrically connected to a low potential power source;

sources of the third transistor and the fifth transistor are eachelectrically connected to a high potential power source; and

an amplitude of a signal inputted to each of gates of the firsttransistor, the second transistor, the fourth transistor, and the fifthtransistor, and to a source of the fourth transistor is smaller than apotential difference between the high potential power source and the lowpotential power source.

According to the present invention, there is provided a shift registerincluding:

a clocked inverter including a first transistor to a third transistorconnected in series; and

a compensation circuit including a fourth transistor and a fifthtransistor connected in series, in which:

sources of the first transistor and the fifth transistor are eachelectrically connected to a first power source;

a source of the third transistor is electrically connected to a secondpower source;

a gate of the first transistor is connected to an output terminal of thecompensation circuit;

a pulse generated at an (n−1)th stage is inputted to an input terminalof the compensation circuit arranged at an n-th stage; and

a pulse or a clock signal generated at an (n−2)th stage is inputted to asource of the fourth transistor arranged at the n-th stage.

The present invention having the structures described above provides aclocked inverter and a shift register that are capable of relaxing aninfluence of fluctuation in the threshold value of a TFT, achieving anoperation without level-shifting a signal having a voltage amplitudethat is smaller than the power source voltage amplitude of a circuit,and performing a high frequency operation and a low voltage operation. ANAND and a NOR having a low input load and a high output capability arealso provided.

Also, no level shifter is arranged in an external circuit, so that theminiaturization of a casing, the reduction of manufacturing costs, andthe reduction of power consumption are realized. Further, the shiftregister is achieved without arranging any level shifter in an internalcircuit. As a result, such problems that the waveform of CK is delayedand blunted and that the voltage of a power source line arranged in theinternal circuit is dropped are solved. Also, the reduction of an areaoccupied by a driver circuit in the internal circuit, the reduction ofpower consumption, and a high frequency operation are realized.

It should be noted here that the clocked inverter is not limited to thetype shown in FIGS. 11A to 11D and includes a type, in which the clockedinverter shown in FIG. 11A is modified and a clock signal is notdirectly inputted, such as a clocked inverter 10 in FIG. 1A, a clockedinverter 10 in FIG. 1C, a clocked inverter 10 in FIG. 2A, a clockedinverter 10 in FIG. 2C, clocked inverters 10 and 17 in FIG. 3A, clockedinverters 10 and 17 in FIG. 3C, and clocked inverters 10 and 17 in FIG.12A

BRIEF DESCRIPTION OF THE DRAWINGS

In the accompanying drawings:

FIGS. 1A to 1D are circuit diagrams of one stage of a shift register andtiming charts;

FIGS. 2A to 2D are circuit diagrams of one stage of a shift register andtiming charts;

FIGS. 3A to 3D are circuit diagrams of one stage of a shift register andtiming charts;

FIGS. 4A to 4D are NAND circuit diagrams and timing charts;

FIGS. 5A to 5D are NOR circuit diagrams and timing charts;

FIGS. 6A and 6B are circuit diagrams of one stage of a shift register;

FIGS. 7A and 7B are circuit diagrams of one stage of a shift register;

FIGS. 8A to 8C show a panel;

FIGS. 9A to 9H show electronic appliances according to the presentinvention;

FIGS. 10A and 10B are a mask layout and a photograph of the top surfacethereof;

FIGS. 11A to 11D are circuit diagrams of a clocked inverter, a NAND anda NOR; and

FIGS. 12A and 12B are a circuit diagram of one stage of a shift registerand a timing chart, respectively.

DESCRIPTION OF THE PREFERRED EMBODIMENTS Embodiment Mode 1

This embodiment mode of the present invention will be described belowwith reference to FIGS. 1A to 1D. In this embodiment mode, as anexample, it is assumed that CK is switched between 5 V (H level) and 2 V(L level), VDD (high potential power source) is 7 V, and VSS (lowpotential power source) is 0 V. That is, it is assumed that theamplitude of CK is 3 V and a power source voltage amplitude is 7 V.

A first structure of the present invention will be described withreference to FIG. 1A FIG. 1A is a circuit diagram showing structuralelements of a shift register arranged in the n-th stage. Each stage isformed by a clocked inverter 10 including TFTs 11 to 13 connected inseries, a compensation circuit 19 a including TFTs 14 a and 15 aconnected in series, an inverter 16, and a clocked inverter 17. Theshift register is formed by cascade-connecting the respective stages, inwhich these circuits are arranged, with signals from CK and CKB beingalternately inputted at the respective stages.

The gate of the TFT 11 is connected to a clock signal line and receivesCK. The gate of the TFT 12 receives a signal S that is a start pulse orthe output of the inverter 16 arranged at the (n−1 )th stage, the gatesof the TFTs 14 a and 15 a receive a signal SB that is the invertedsignal of the signal S, and the source of the TFT. 14 a receives theoutput of the clocked inverter 10 arranged at the (n−2 )th stage. Notethat in the drawings, the output of the clocked inverter 10 arranged atthe (n−2 )th stage is denoted as the “two-stage-before signal”.

In the present invention, in the compensation circuit 19, the gates ofthe TFTs 14 a and 15 a connected to each other are each set as an inputterminal, and the drains of the TFT's 14 a and 15 a connected to eachother are each set as an output terminal.

Operations will be described by following a timing chart shown in FIG.1B. In FIG. 1B, one half of the cycle of the clock signal is set as “T”.Operations in periods T1 and T2 will be described below.

In the period T1, the two-stage-before signal is at VSS, the signal S isat VDD, the signal SB is at VSS, and CK is at the H level (5 V), so thatthe TFT 12 is turned off, the TFT 14 a is turned off, the TFT 15 a isturned off, and the TFT 13 is turned off. In this case, VDD is held by aloop formed by the inverter 16 and the clocked inverter 17 and an outputOUT assumes VDD.

Following this, when time advances from the period T1 to the period T2,the two-stage-before signal is switched from VSS to VDD, the signal Sremains at VDD, the signal SB remains at VSS, and CK is switched to theL level (2 V), so that the TFT 12 remains turned off, the TFT 14 a isturned on, and the TFT 15 a remains turned off. In this case, the signalinputted to the gate of the TFT 13 is switched to VDD, so that the TFT13 is switched from an OFF state to an ON state. As a result, the outputOUT assumes VSS. In the present invention, the switching of OUT from VDDto VSS is referred to as the “falling”.

Next, a second structure of the present invention will be described withreference to FIG. 1C. FIG. 1C is a circuit diagram showing structuralelements of a shift register arranged at the nth stage. The differencesfrom the first structure described above are that a compensation circuit19 b including TFTs 14 b and 15 b connected in series is connected tothe gate of the TFT 11, the P-type TFT 12 is eliminated and an N-typeTFT 18 is arranged instead, the source of the TFT 15 b receives theoutput of the clocked inverter 10 arranged at the (n−2 )th stage, thegate of the TFT 18 receives the signal S, and the clock signal line isconnected to the gate of the TFT 13 and CK is inputted to the gate ofthe TFT 13.

Next, operations in periods T1 and T2 will be described with referenceto a timing chart shown in FIG. 1D. Note that the operation according tothe second structure is similar to the operation according to the firststructure described above and therefore will be described in brief.

In the period T1, the output OUT assumes VSS. When time advances fromthe period T1 to the period T2, the two-stage-before signal inputted tothe gate of the TFT 11 is switched from VDD to VSS, so that the TFT 11is turned on. On the other hand, the TFT 18 is turned off, so that theoutput OUT assumes VDD. In the present invention, the switching of OUTfrom VSS to VDD is referred to as the “rising”.

The present invention having the first structure described above is veryeffective for the falling, and the present invention having the secondstructure described above is very effective for the rising. As a result,the following effect (1) is provided.

The effect (1) will first be described. When CK is inputted as it is tothe source of the TFT 14 a shown in FIG. 1A or the source of the TFT 15b shown in FIG. 1C, there arises a problem that the TFT described aboveis turned on earlier than a desired timing because the amplitude of CKis small. In more detail, there arises a problem that a signal having adotted waveform 170 in FIG. 1B or a signal having a dotted waveform 171in FIG. 1D is generated. That is, there arises a problem that when aleak current is large, shift of pulse does not occur. In the presentinvention, however, the two-stage-before signal is used, so that it ispossible to turn on the TFT described above at desired timing withoutbeing turned on too early. As a result, it is possible to solve such theproblem that the shift of the pulse does not occur.

In addition to the effect (1) described above, the present inventionhaving the first structure or the second structure described aboveprovides the following advantageous effects (2) and (3).

First, the effect (2) will be described. In usual cases, a clockedinverter is formed by four TFTs that are two N-type TFTs connected inseries and two P-type TFTs connected in series. Also, in order to obtaina large on-current, the gate widths (W) of the two TFTs connected inseries are set large, which results in the necessity to increase thegate width of a TFT whose gate functions as a load. As a result, theoverall load is increased and a high frequency operation is obstructed.In the present invention, however, it is possible to change adouble-gate TFT (two TFTs connected in series) into a single-gate TFT.In the case of the structure shown in FIG. 1A, for instance, it hasconventionally been required to arrange two N-type TFTs connected inseries. It is, however, sufficient that only one N-type TFT 13 isarranged in the present invention. As a result, in the presentinvention, it is not required to increase the gate widths of the TFTsand it is possible to reduce the sizes of the TFTs, which makes itpossible to realize a high integration. Further, the burden on anelement, whose gate (gate capacitance) functions as a load, is reducedand the overall load is also reduced, so that a high frequency operationbecomes possible.

Next, the effect (3) will be described. Two TFTs of the same conductivetype connected in series are weak in current performance (power). In thepresent invention, however, it is possible to change a double-gate TFTinto a single-gate TFT, which makes it possible to enhance the currentperformance of the TFT. In the structure shown in FIG. 1A, for instance,it is possible to enhance the current performance of the N-type TFT 13.Also, in the structure shown in FIG. 1C, it is possible to enhance thecurrent performance of the P-type TFT 11. Note that the currentperformance is defined as K=μC_(ox)W/2L, where K is current performance,μ is mobility of carrier, C_(ox) is capacitance of gate insulating filmper unit area, W is channel width, and L is channel length.

As described above, the structure shown in FIG. 1A is very effective forthe falling and rising. In FIGS. 1A and 1B, however, when time advancesto the period T3, S is switched to VSS, SB is switched to VDD, and CK isswitched to the H level, so that the TFT 12 is turned on, the TFT 13 isturned off, and the TFT 11 is turned on or off depending on itsthreshold value. If the threshold value of the TFT 11 is lower than adesired value, there arises a case in which the TFT 11 is turned on andtherefore the shift register does not operate properly.

In view of this problem, a structure that is effective for the holdingVSS without making OUT rising earlier in the period T3 will be proposedas a third structure of the present invention.

The third structure of the present invention will be described withreference to FIG. 2A. FIG. 2A is a circuit diagram showing structuralelements of a shift register arranged in the n-th stage. Each stage isformed by a clocked inverter 10 including TFTs 11 and 13 connected inseries, a compensation circuit 19 a including TFTs 14 a and 15 aconnected in series, a compensation circuit 19 b including TFTs 14 b and15 b, an inverter 16, and a clocked inverter 17 including TFTs 22 to 25.The shift register is formed by cascade-connecting the respectivestages, in which these circuits are arranged, with CK and CKB beingalternately inputted at the respective stages. The differences betweenthe structure shown in FIG. 2A and the structure shown in FIG. 1A residein that the TFT 12 is eliminated, the output of the compensation circuit19 b is connected to the gate of the TFT 11, SB is connected to theinput of the compensation circuit 19 b, VDD is connected to the sourceof the TFT 14 b, CK is connected to the source of the TFT 115 b, and thechannel width is so set large that the current performance of the TFT 24and the TFT 25 are enhanced.

Operations of a structure shown in FIG. 2A in periods T1 and T2 will bedescribed with reference to a timing chart shown in FIG. 2B.

In the period T1, the two-stage-before signal is at VDD, the signal SBis at VSS, and the clock signal CK is at the L level, so that the TFT 14a is turned on, the TFT 15 a is turned off, the TFT 13 is turned on, theTFT 14 b is turned on, the TFT 15 b is turned off, and the TFT 11 isturned off. As a result, the output OUT assumes VSS.

Next, in the period T2, the two-stage-before signal remains at VDD, thesignal SB is switched to VDD, and the clock signal CK is switched to theH level, so that the TFT 13 is turned off and the TFT 11 is turned on oroff. Under this state, OUT at VSS is held by a loop formed by theinverter 16 and the clocked inverter 17, and VSS is continuouslyoutputted as OUT. Note that in the present invention, the operationperformed in the period T2 is referred to as the “holding”. Thisstructure is very effective for the holding. The holding operation inthe period T2 will be described in more detail below.

In the period T2, the signal SB is at VDD (7 V). VGS of the TFT 15 bbecomes 2 V when the signal SB is at VDD (7 V) and CK is at the H level(5 V).

Under this condition, if the threshold voltage (|VTH|) of the TFT 15 bis equal to or less than 2 V, the TFT 15 b is turned on and CK (H level,5 V) is inputted to the gate of the TFT 11. Then, the TFT 11 is turnedon or off depending on its threshold voltage.

If the TFT 11 is turned on, it attempts to output VDD as OUT. However,the current capacities of the TFT 24 and the TFT 25 of the clockedinverter 17 holding VSS are set large, so that VSS is outputted and atheoretically proper operation is performed. As a result, as indicatedby a dotted waveform 172 in the timing chart shown in FIG. 2B, such asituation is prevented, in which a signal outputted as OUT is notcorrectly held and switching from VSS to VDD is performed earlier thandesired timing.

Also, even if a correct operation is performed as described above, whenthe P-type TFT 11 that should be turned off remains turned on, therearises a problem that a leak current flows between VDD and VSS andtherefore the current consumption is increased. In such a case, as shownin FIG. 2A, inverters 20 and 21 may be connected to the gates of the TFT14 b and the TFT 15 b. With this structure, as indicated by a dottedwaveform 174 in FIG. 2B, it is possible to delay the signal SB and todelay the timing at which the TFT 15 b is turned on, which makes itpossible to delay timing at which a leak current flows. Note that thenumber of inverters to be connected is not specifically limited so longas no theoretical difference occurs, although the degree of the delay isset at equal to or less than one half of the cycle of CK.

On the other hand, if the threshold voltage (|VTH|) of the TFT 11 or theTFT 15 b is equal to or more than 2 V, the TFT 15 b is not turned on andno leak current is generated. If it is possible to prevent thegeneration of the leak current, an increase in current consumption andthe rising of the waveform of the output signal OUT earlier than thedesired timing are prevented. As a result, a signal having a stabilizedwaveform is generated.

Also, in the period T3 in FIGS. 1C and 1D, there arises a case in whichthe threshold value of the N-type TFT 15 b is lower than a desired valueand the N-type TFT 15 b is turned on. In this case, it is impossible tohold OUT at VDD and the shift register does not operate properly.

In view of this, a structure that is effective for the holding of OUT atVDD in the period T3 will be proposed as a fourth structure of thepresent invention.

A fourth structure of the present invention will be described withreference to FIG. 2C. FIG. 2C is a circuit diagram showing structuralelements of a shift register arranged in the n-th stage. The differencesof the structure shown in FIG. 2C from the second structure reside inthat the TFT 18 is eliminated, the output of the compensation circuit 19a is connected to the gate of the TFT 13, SB is connected to the inputof the compensation circuit 19 a, CK is connected to the source of theTFT 14 a, VSS is connected to the source of the TFT 15 a, and thechannel width is set large so that the current performance of the TFT 22and the TFT 23 are enhanced.

Next, operations in periods T1 and T2 will be described by following atiming chart shown in FIG. 2D. Note that the operation according to thestructure shown in FIG. 2C is similar to the operation according to thestructure shown in FIG. 2A described above and therefore will bedescribed in brief.

In the period T1, the two-stage-before signal is at VSS, the signal SBis at VDD, and the clock signal CK is at the H level, so that the TFT 14b is turned off, the TFT 15 b is turned on, and the TFT 11 is turned on.As a result, the output OUT assumes VDD.

Next, in the period T2, the two-stage-before signal remains at VSS, thesignal SB is switched to VSS, and the clock signal CK is switched to theL level, so that the TFT 11 is turned off and the TFT 13 is turned on oroff. Under this state, OUT at VDD is held by a loop formed by theinverter 16 and the clocked inverter 17, and VDD is continuouslyoutputted as OUT. This structure is very effective for the holding. Theoperation in the period T2 will be described in more detail below.

In the period T2, the signal SB is at VSS (0 V). VGS of the TFT 14 abecomes 2 V when the signal SB is at VSS (0 V) and CK is at the L level(2 V).

Under this condition, if the threshold voltage (|VTH|) of the TFT 14 ais equal to or less than 2 V, the TFT 14 a is turned on and CK (L level,2 V) is inputted to the gate of the TFT 13. Then, the TFT 13 is turnedon or off depending on its threshold voltage.

If the TFT 13 is turned on, it attempts to output VSS as OUT. However,the current capacities of the TFT 22 and the TFT 23 of the clockedinverter 17 holding VDD are set large, so that a theoretically properoperation is performed. As a result, as indicated by a dotted waveform173 in the timing chart shown in FIG. 2D, a situation is prevented, inwhich a signal outputted as OUT is not correctly held and switching fromVDD to VSS is performed earlier than a desired timing.

Also, even if a correct operation is performed as described above, whenthe N-type TFT 13 that should be turned off remains turned on, therearises a problem that a leak current flows between VDD and VSS andtherefore the current consumption is increased. In such a case, as shownin FIG. 2C, inverters 20 and 21 may be connected to the gates of the TFT14 a and the TFT 15 a. With this structure, as indicated by a dottedwaveform 175 in FIG. 2D, it is possible to delay the signal SB and todelay the timing at which the P-type TFT 14 a is turned on, which makesit possible to delay a timing at which a leak current flows. Note thatthe number of inverters to be connected is not specifically limited solong as no theoretical difference occurs, although the degree of thedelay is set at equal to or less than one half of the cycle of CK.

On the other hand, if the threshold voltage (|VTH|) of the TFT 13 or theTFT 14 a is equal to or more than 2 V, the TFT 13 is not turned on andno leak current is generated. If it is possible to prevent thegeneration of the leak current, an increase in current consumption isprevented. Also, the waveform of the output signal OUT is not turned onearlier than the desired timing. As a result, a signal having astabilized waveform is generated.

In conclusion, the present invention having the third or fourthstructure described above is very effective for the holding and providesthe following effects (4) and (5).

First, the effect (4) will be described. When the threshold voltage(|VTH|) of the TFT 15 b in the structure shown in FIG. 2A or thethreshold voltage (|VTH|) of the TFT 14 a in the structure shown in FIG.2C is equal to or less than a desired value (2 V), multiple invertersmay be connected to the input terminal of the compensation circuit 19 aor 19 b. With this structure, even if the threshold voltage of the TFTdescribed above is equal to or less than the desired value, it ispossible to delay the timing at which the leak current is generated.

Next, the effect (5) will be described. Conventionally, there has been aproblem that a TFT that should be turned off remains turned on and aleak current flows between VDD and VSS, resulting in the increase of thecurrent consumption. In the structure shown in FIG. 2A, for instance,the P-type TFT 11 that should be turned off remains turned on. Also, inthe structure shown in FIG. 2C, the N-type TFT 13 that should be turnedoff remains turned on. In the present invention, however, when thethreshold voltage (|VTH|) of the TFT 11 or the TFT 15 b in the structureshown in FIG. 2A or the threshold voltage (|VTH|) of the TFT 13 or theTFT 14 a in the structure shown in FIG. 2C is equal to or more than thedesired value (2 V), it is possible to suppress the generation of theleak current.

Also, as in the case of the first and second structures, the presentinvention having the third or fourth structure described above providesthe advantageous effects (2) and (3) described above.

In the structure shown in FIGS. 2A and 2B, however, in order to performa theoretically proper operation even if the TFT 11 is turned on, thecurrent capacities of the TFTs 24 and 25 in the holding clocked inverterare set large. Therefore, there occurs a case in which even when timeadvances from the period T2 to the period T3 and CK is switched to the Llevel, the OUT is not switched to VDD and the shift register does notoperate properly.

In view of this, a structure that is capable of obtaining a stabilizedwaveform of OUT in the holding period and is effective for the risingfrom the period T2 to the period T3 will be proposed as a fifthstructure of the present invention.

The fifth structure of the present invention will be described withreference to FIG. 3A FIG. 3A is a circuit diagram showing structuralelements of a shift register arranged at the n-th stage. Each stage isformed by a clocked inverter 10 including TFTs 11 and 13 connected inseries, a compensation circuit 19 a including TFTs 14 a and 15 a, acompensation circuit 19 b including TFTs 14 b and 15 b, an inverter 16,a clocked inverter 17 including TFTs 22 to 24 connected in series, and acompensation circuit 19 c including an N-type TFT 34 and an analogswitch 35. A shift register is formed by cascade-connecting therespective stages, in which these circuits are arranged, with CK and CKBbeing alternately inputted at the respective stages. The differencesfrom FIG. 2A reside in that the TFT 25 is eliminated from the holdingclocked inverter 17, the output from the compensation circuit 19 c isconnected to the gate of the TFT 24, the input terminal of the inverter16 (that is, the output terminal of the clocked inverter 10) isconnected to the gate of the TFT 34 of the compensation circuit 19 c andto the gate on the P-type TFT side of the analog switch 35, the outputof the inverter 16 is connected to the gate on the N-type TFT side ofthe analog switch 35, VSS is connected to the source of the TFT 34, andCK is connected to the source of the analog switch 35.

The gate of the TFT 22 is connected to a clock bar signal line andreceives CKB, and the gate of the TFT 23 receives the output of theinverter 16. Also, the current performance of the TFT 24 is set large.In more detail, if it is assumed that “W₂₄/L:W₁₁/L=x:y”, W₂₄/L of theTFT 24 and W₁₁/L of the TFT 11 are respectively set as “y=1, x≧1” (whereW is a gate width and L is a gate length).

Operations in periods T1 to T3 will be described by following a timingchart shown in FIG. 3B. In the period T1, VSS is outputted from theclocked inverter 10.

Next, the operation in the period T2 will be described. In the clockedinverter 17, CKB (L level, 2 V) is inputted to the gate of the TFT 22and the TFT 22 is turned on. The inverted signal (VDD) of OUT isinputted to the gate of the TFT 23 and the TFT 23 is turned off. Theoutput OUT (VSS) is inputted to the gate of the TFT 34 and the TFT 34 isturned off. The signal CK (H level, 5 V) is inputted to the gate of theTFT 24 via the analog switch 35 and the TFT 24 is turned on. Under thiscondition, the TFT 23 is turned off and the TFT 24 is turned on, so thatVSS is outputted.

Also, in the clocked inverter 10, the TFT 11 is turned on or off. Evenif the TFT 11 is turned on, the current performance of the TFT 24 is setlarge, so that VSS is outputted with stability in the period T2.

It is desired that when time advances from the period T2 to the periodT3, the output of the clocked inverter 10 be switched from VSS to VDDwith precision. However, the current performance of the N-type TFT 24 isset large, so that as indicated by a waveform 176 in the timing chartshown in FIG. 3B, there arises a case where it is impossible to performthe switching from VSS to VDD and the shift register does not operateproperly. In the present invention, however, the followingcountermeasures are taken in order to prevent such a situation.

When the time advances from the period T2 to the period T3, the clockedinverter 10 attempts to switch its output from VSS (0 V) to VDD (7 V).However, the current performance of the N-type TFT 24 possessed by theclocked inverter 17 is set large, so that such a case arises, in whichalthough |VGS| applied to the TFT 11 changes from 2 V to 5 V and anattempt is made to output VDD as OUT, it is impossible to increase theoutput from 0 V to 7 V. In this case, the output of the inverter 16 doesnot become 0 V, and 7 V is continuously inputted to the holding clockedinverter 17. As a result, the on/off states of the TFT 23 and the TFT 24are not interchanged and VSS (0 V) is continuously outputted as OUT,which means that the shift register does not operate properly.

In the present invention, however, even if the output of the clockedinverter 10 is not switched from VSS (0 V) to VDD (7 V), if the outputOUT changes by a degree at least equal to the threshold value of the TFT34 at the moment of changing of VGS applied to the TFT 11 from 2 V to 5V, the TFT 34 is turned on and the TFT 24 is compulsively turned off. Asa result, it is possible for the TFT 11 to raise the output OUT to VDDwithout being influenced by the TFT 24. In addition, the rising of OUTcan be performed at desired timing. Also, when the TFT 35 is replacedwith an analog switch, the L level of CK is inputted to the gate of theTFT 24 at this point of time. If the threshold value of the TFT 24 isequal to or more than 2 V, the TFT 24 is turned off. Also, even if thethreshold value is equal to or less than 2 V and the TFT 24 is turnedon, |VGS| is reduced from 5 V to 2 V, so that the holding ability isweakened. As a result, the output OUT changes easily.

The current performance of the TFT 24 is also ascribable to thethreshold value. Therefore, it is conceivable that when the thresholdvalue of the N-type TFT is lowered and the current performance of theTFT 24 is enhanced, the threshold value of the TFT 34 having the samepolarity is lowered. As a result, the turning-on is performed even ifthe changing degree of OUT is small. In contrast to this, even if thethreshold value of the TFT 34 is high, the threshold value of the TFT 24is also high in this case and the holding ability is weakened. As aresult, a proper operation is performed without any problems.

In conclusion, the present invention having the fifth structuredescribed above is very effective for the holding and rising andprovides the following effects (6) and (7).

First, the effect (6) will be described. In the present invention, thecurrent performance of the N-type TFT 24 possessed by the clockedinverter 17 is set large. When VSS is held by a loop formed by theinverter 16 and the clocked inverter 17, the current performance of theTFT 24 is set large, so that it is possible to output VSS withstability.

Next, the effect (7) will be described. At the rising of the output ofthe clocked inverter 10 from VSS to VDD, the current performance of theN-type TFT 24 possessed by the clocked inverter 17 is set large, so thatthere arises a case in which the rising is not performed and a properoperation is not performed. However, the timing of this rising isdetermined by the P-type TFT 11 possessed by the clocked inverter 10. Ifthe output OUT changes at the moment of changing of VGS of the TFT 11,the N-type TFT 34 is turned on at the time when its threshold value isexceeded. As a result, the output OUT rises with precision.

Similarly in FIGS. 2C and 2D, there arises a case in which even whentime advances from the period T2 to the period T3 and CK is switched tothe H level, the OUT is not switched to VSS and the shift register doesnot operate properly.

In view of this, a structure that is capable of obtaining a stabilizedwaveform of OUT in the holding period and is effective for the risingfrom the period T2 to the period 13 will be proposed as a sixthstructure of the present invention.

The sixth structure of the present invention will be described withreference to FIG. 3C. FIG. 3C is a circuit diagram showing structuralelements of a shift register arranged at the n-th stage. Each stage isformed by a clocked inverter 10 including TFTs 11 and 13 connected inseries, a compensation circuit 19 a including TFTs 14 a and 15 a, acompensation circuit 19 b including TFTs 14 b and 15 b, an inverter 16,a clocked inverter 17 including TFTs 23 to 25 connected in series, and acompensation circuit 19 d including an P-type TFT 37 and an analogswitch 35. A shift register is formed by cascade-connecting therespective stages, in which these circuits are arranged, with CK and CKBbeing alternately inputted at the respective stages. The differencesfrom FIG. 2C are that the TFT 22 is eliminated from the holding clockedinverter 17, the output from the compensation circuit 19 d is connectedto the gate of the TFT 23, the input terminal of the inverter 16 (thatis, the output terminal of the clocked inverter 10) is connected to thegate of the P-type TFT 37 of the compensation circuit 19 d and to thegate on the N-type TFT side of the analog switch 35, the output of theinverter 16 is connected to the gate on the P-type TFT side of theanalog switch 35, VDD is connected to the source of the TFT 37, and CKis connected to the source of the analog switch 35.

The gate of the TFT 25 is connected to a clock bar signal line andreceives CK, and the gate of the TFT 37 receives the output (OUT) of theclocked inverter 10. Also, the current performance of the TFT 23 is setlarge. In more detail, if it is assumed that “W₂₃/L:W₁₃/L=x:y”, W₂₃/L ofthe TFT 23 and W₁₃/L of the TFT 13 are respectively set as “y=1, x≧1”(where W is a gate width and L is a gate length).

Operations in periods T1 to T3 will be described with reference to atiming chart shown in FIG. 3D. In the period T1, VDD is outputted fromthe clocked inverter 10.

Next, an operation in the period T2 will be described. In the clockedinverter 17, CKB (H level, 5 V) is inputted to the gate of the TFT 25and the TFT 25 is turned on. The inverted signal (VSS) of OUT isinputted to the gate of the TFT 24 and the TFT 24 is turned off. Theoutput OUT (VDD) is inputted to the gate of the TFT 37 and the TFT 37 isturned off. The signal CK (L level, 2 V) is inputted to the gate of theTFT 23 via the analog switch 35 and the TFT 23 is turned on. Under thiscondition, the TFT 24 is turned off and the TFT 23 is turned on, so thatVDD is outputted.

Also, in the clocked inverter 10, the TFT 13 is turned on or off. Evenif the TFT 13 is turned on, the current performance of the TFT 23 is setlarge, so that VDD is outputted with stability in the period T2.

It is desired that when time advances from the period T2 to the periodT3, the output of the clocked inverter 10 be switched from VDD to VSSwith precision. However, the current performance of the P-type TFT 23 isset large, so that as indicated by a waveform 177 in the timing chartshown in FIG. 3D, there arises a case in which it is impossible toperform the switching from VDD to VSS and the shift register does notoperate properly. In the present invention, however, the followingcountermeasures are taken in order to prevent such a situation.

When time advances from the period T2 to the period T3, the clockedinverter 10 attempts to switch its output from VDD (7 V) to VSS (0 V).However, the current performance of the P-type TFT 23 possessed by theclocked inverter 17 is set large, so that there occurs a case in whichalthough VGS applied to the TFT 13 changes from 2 V to 5 V and anattempt is made to output VSS as OUT, it is impossible to decrease theoutput from 7 V to 0 V. In this case, the output of the inverter 16 doesnot become 7 V, and 0 V is continuously inputted to the holding clockedinverter 17. As a result, the on/off states of the TFT 23 and the TFT 24are not interchanged and VDD (7 V) is continuously outputted as OUT,which means that the shift register does not operate properly.

In the present invention, however, even if the output of the clockedinverter 10 is not switched from VDD (7 V) to VSS (0 V), if the outputOUT changes by a degree at least equal to the threshold value of the TFT37 at the moment of changing of VGS applied to the TFT 13 from 2 V to 5V, the TFT 37 is turned on and the TFT 23 is compulsively turned off. Asa result, it is possible for the TFT 13 to lower the output OUT to VSSwithout being influenced by the TFT 23. In addition, the falling of OUTis performed at desired timing. Also, when the TFT 35 is replaced withan analog switch, the H level of CK is inputted to the gate of the TFT23 at this point. If the threshold value of the TFT 23 is equal to ormore than 2 V, the TFT 23 is turned off. Also, even if the thresholdvalue is less than 2 V and the TFT 24 is turned on, |VGS| is reducedfrom 5 V to 2 V, so that the holding ability is weakened. As a result,the output OUT changes easily.

The current performance of the TFT 23 is also ascribable to thethreshold value. Therefore, it is conceivable that when the thresholdvalue of the P-type TFT is lowered and the current performance of theTFT 23 is enhanced, the threshold value of the TFT 37 having the samepolarity is lowered. As a result, the turning-on is performed even ifthe changing degree of OUT is small. In contrast to this, even if thethreshold value of the TFT 37 is large, the threshold value of the TFT23 is also large in this case and the holding ability is weakened. As aresult, a proper operation is performed without any problems.

In conclusion, the present invention having the sixth structuredescribed above is very effective for the holding and the falling, andprovides the following effects (8) and (9).

First, the effect (8) will be described. In the present invention, thecurrent performance of the P-type TFT 23 possessed by the clockedinverter 17 is set large. When VDD is held by a loop formed by theinverter 16 and the clocked inverter 17, the current performance of theTFT 23 is set large, so that it is possible to output VDD withstability.

Next, the effect (9) will be described. At the falling of the output ofthe clocked inverter 10 from VDD to VSS, the current performance of theP-type TFT 23 possessed by the clocked inverter 17 is set large, so thatthere occurs a case in which the falling is not caused and a properoperation is not performed. However, the timing of this falling isdetermined by the N-type TFT 13 possessed by the clocked inverter 10. Ifthe output OUT changes at the moment of changing of VGS of the TFT 13,the P-type TFT 37 is turned on at the time when its threshold value isexceeded. As a result, the output OUT is lowered with precision.

Embodiment Mode 2

It is possible to use the first to sixth structures described above withreference to FIGS. 1A to 1D, 2A to 2D, and 3A to 3D by freely combiningthem. In this embodiment mode, an example of the combination will bedescribed with reference to FIGS. 6A and 6B and FIGS. 7A and 7B. Notethat in those drawings, a signal S is a start pulse or an output of aclocked inverter 16 arranged at the (n−1 )th stage, and a signal SBcorresponds to the inverted signal of the signal S. Also, the term“two-stage-before signal” corresponds to the output of the clockedinverter 10 arranged at the (n−2 )th stage.

FIG. 6A is a circuit diagram in which the third structure (see FIG. 2A)and the fifth structure (see FIG. 3A) are combined, and shows structuralelements of a shift register arranged at the n-th stage. Each stage isformed by a clocked inverter 10 including TFTs 71 to 73 connected inseries, an inverter 16, a clocked inverter 17 including TFTs 74 and 75connected in series, TFTs 76 and 77 connected in series, inverters 78and 79, a TFT 80, and an analog switch 81. The shift register is formedby cascade-connecting the respective stages, in which these circuits arearranged, with CK and CKB being alternately inputted at the respectivestages.

FIG. 6B is a circuit diagram in which the second structure (see FIG.1C), the fourth structure (see FIG. 2C), and the sixth structure (seeFIG. 3C) are combined with each other, and FIG. 6B shows structuralelements of a shift register arranged at the n-th stage. Each stage isformed by a clocked inverter 10 including TFTs 91 to 93 connected inseries, an inverter 16, a clocked inverter 17 including TFTs 94 and 95connected in series, TFTs 96 and 97 connected in series, TFTs 98 and 99connected in series, inverters 120 and 121, a P-type TFT 122, and ananalog switch 123. The shift register is formed by cascade-connectingrespective stages, in which these circuits are arranged, with CK and CKBbeing alternately inputted at the respective stages.

FIG. 7A is a circuit diagram in which the fourth structure (see FIG. 2C)and the sixth structure (see FIG. 3C) are combined with each other, andFIG. 7A shows structural elements of a shift register arranged at then-th stage. Each stage is formed by a clocked inverter 10 including TFTs131 to 133 connected in series, an inverter 16, a clocked inverter 17including TFTs 134 and 135 connected in series, TFTs 136 and 137connected in series, inverters 138 and 139, a P-type TFT 140, and ananalog switch 141. The shift register is formed by cascade-connectingthe respective stages, in which these circuits are arranged, with CK andCKB being alternately inputted at the respective stages.

FIG. 7B is a circuit diagram in which the first structure (see FIG. 1A),the third structure (see FIG. 2A) and the fifth structure (see FIG. 3A)are combined with each other, and FIG. 7B shows structural elements of ashift register arranged at the n-th stage. Each stage is formed by aclocked inverter 10 including TFTs 151 to 153 connected in series, aninverter 16, a clocked inverter 17 including TFTs 154 and 155 connectedin series, TFTs 156 and 157 connected in series, TFTs 158 and 159connected in series, inverters 160 and 161, an N-type TFT 162, and ananalog switch 163. The shift register is formed by cascade-connectingthe respective stages, in which these circuits are arranged, with CK andCKB being alternately inputted at the respective stages.

It should be noted here that when some or all of the first to sixthstructures described above are combined and used, unnecessary TFTs maybe eliminated if the circuit operates without any troubles. In thestructure shown in FIGS. 6A and 7B, the TFT 22 in FIG. 3A is indeedeliminated. Also, in the structure shown in FIGS. 6B and 7A, the TFT 25shown in FIG. 3C is indeed eliminated. In a like manner, TFTs may beadditionally arranged as necessary if no trouble occurs in itsoperation.

Embodiment Mode 3

This embodiment mode according to the present invention will bedescribed with reference to FIGS. 10A and 10B.

FIG. 10A shows a plan layout view (top view) of the circuit diagramshown in FIG. 6B. FIG. 10B shows a photograph of a panel that isactually made, magnified by a light microscope.

Reference numerals and symbols in FIGS. 10A and 10B correspond to thosein FIG. 6B, and thus the description is omitted here. In FIGS. 10A and10B, a P-type TFT 16 a and an N-type TFT 16 b constitute an inverter 16,and a P-type TFT 123 a and an N-type TFT 123 b constitute an analogswitch 123.

The W (gate width) of the TFT 94 is set large. If another TFT that isconnected with the TFT 94 in series and has the same size as the TFT 94is required, the layout area becomes larger. However, only one TFT 94whose W is set large is required in the present invention, and thereforethe expansion of the layout area is suppressed.

Embodiment Mode 4

An embodiment mode of the present invention that is different from theabove embodiment modes will be described with reference to FIGS. 4A to4D and FIGS. 5A to 5D.

A NAND of the present invention will be described with reference toFIGS. 4A to 4D. FIG. 4A is a circuit diagram of the NAND that includesP-type TFTs 51 and 52 connected in parallel, an N-type TFT 54, and acompensation circuit 19 including a P-type TFT 55 and an N-type TFT 56connected in series. The gate of the TFT 51 receives Vin1, the gate ofthe TFT 52 and the source of the TFT 55 receive Vin2, and the gates ofthe TFTs 55 and 56 receive VinB1 that is the inverted signal of Vin1.

How this NAND operates will be described by following a timing chartshown in FIG. 4B. In the period T1, Vin1 is at the H level, VinB1 is atthe L level, and Vin2 is at the L level, so that the TFT 51 is turnedoff, the TFT 52 is turned on, the TFT 55 is turned on, and the TFT 56 isturned off. Also, Vin2 (at the L level) is inputted to the TFT 54 viathe TFT 55, so that the TFT 54 is turned off. As a result, the outputOUT assumes VDD. In the period T2, Vin1 remain at the H level, VinB1remains at the L level, and Vin2 is switched to the H level, so that theTFT 51 remains turned off, the TFT 52 is turned off, the TFT 55 remainsturned on, and the TFT 56 remains turned off. Also, VinB1 (at the Llevel) is inputted to the TFT 54 via the TFT 55, so that the TFT 54 isturned on. As a result, the output OUT assumes VSS.

In the period T3, Vin1 is switched to the L level, VinB1 is switched tothe H level, and Vin2 remains at the H level, so that the TFT 51 isturned on, the TFT 52 remains turned off, the TFT 55 is turned off, andthe TFT 56 is turned on. Also, VSS is inputted to the TFT 54 via the TFT56, so that the TFT 54 is turned off. As a result, the output OUTassumes VDD. In the period T4, Vin1 remains at the L level, VinB1remains at the H level, and Vin2 is switched to the L level, so that theTFT 51 remains turned on, the TFT 52 is turned on, the TFT 55 remainsturned off, and the TFT 56 remains turned on. Also, VSS is inputted tothe TFT 54 via the TFT 56, so that the TFT 54 remains turned off. As aresult, the output OUT assumes VDD.

Next, a structure in which an analog switch 57 is arranged in place ofthe TFT 55 in the structure described above is shown in FIG. 4C. Thestructure shown in FIG. 4C operates by following a timing chart shown inFIG. 4D. Note that the structure shown in FIG. 4C and the operationthereof is similar to the structure shown in FIG. 4A and the operationthereof described above, and therefore will not be described here.

Next, a NOR of the present invention will be described with reference toFIGS. 5A to 5D. FIG. 5A is a circuit diagram of the NOR which includesN-type TFTs 61 and 62 connected in parallel, a P-type TFT 64, and acompensation circuit 19 including a P-type TFT 65 and an N-type TFT 66connected in series. The gate of the TFT 61 receives Vin1, the gate ofthe TFT 62 and the source of the TFT 66 receive Vin2, and the gates ofthe TFTs 65 and 66 receive Vin1 that is the inverted signal of Vin1.

How the NOR operates will be described with reference to a timing chartshown in FIG. 5B. In the period T1, Vin1 is at the L level, VinB1 is atthe H level, and Vin2 is at the H level, so that the TFT 61 is turnedoff, the TFT 62 is turned on, the TFT 65 is turned off, and the TFT 66is turned on. Also, Vin2 (at the H level) is inputted to the TFT 64 viathe TFT 66, so that the TFT 64 is turned off. As a result, the outputOUT assumes VSS. In the period T2, Vin1 remains at the L level, VinB1remains at the H level, and Vin2 is switched to the L level, so that theTFT 61 remains turned off, the TFT 62 is turned off, the TFT 65 remainsturned off, and the TFT 66 remains turned on. Also, Vin2 (at the Llevel) is inputted to the TFT 64 via the TFT 66, so that the TFT 64 isturned on. As a result, the output OUT assumes VDD.

In the period T3, Vin1 is switched to the H level, VinB1 is switched tothe L level, and Vin2 remains at the L level, so that the TFT 61 isturned on, the TFT 62 remains turned off, the TFT 65 is turned on, andthe TFT 66 is turned off. Also, VDD is inputted to the TFT 64 via theTFT 65, so that the TFT 64 is turned off. As a result, the output OUTassumes VSS. In the period T4, Vin1 remains at the H level, VinB1remains at the L level, and Vin2 is switched to the H level, so that theTFT 61 remains turned on, the TFT 62 is turned on, the TFT 65 remainsturned on, and the TFT 66 remains turned off. Also, VDD is inputted tothe TFT 64 via the TFT 65, so that the TFT 64 remains turned off. As aresult, the output OUT assumes VSS.

Next, a structure in which an analog switch 67 is arranged in place ofthe TFT 66 in the structure described above is shown in FIG. 5C. Thestructure shown in FIG. 5C operates by following a timing chart shown inFIG. 5D. Note that the structure shown in FIG. 5C and the operationthereof are similar to the structure shown in FIG. 5A and the operationthereof described above, and therefore are not described here.

The NAND of the present invention having the structure shown in FIG. 4Aor 4C described above and the NOR of the present invention having thestructure shown in FIG. 5A or 5C described above provide the followingadvantageous effect (10).

The effect (10) will be described. In usual cases, the NAND and NOR areeach formed by four TFTs that are two N-type TFTs connected in seriesand two P-type TFTs connected in series. Also, in order to obtain alarge on-current, the gate widths (W) of the two TFTs connected inseries are set large. As a result, it is required to increase the gatewidth of the TFT whose gate functions as a load, which increases theoverall load and obstructs a high frequency operation. In the presentinvention, however, a double-gate TFT (two TFTs connected in series) ischanged into a single-gate TFT. In the structure shown in FIG. 4A, forinstance, it has conventionally been required to arrange two N-type TFTsconnected in series. Only one N-type TFT 13, however, is arranged in thepresent invention. As a result, in the present invention, it is notrequired to increase the gate width of the TFT and it is possible toreduce the size of the TFT, which makes it possible to realize a highintegration. Further, the burden on an element, whose gate (gatecapacitance) functions as a load, is reduced and therefore the overallload is also reduced. As a result, a high frequency operation becomespossible.

In this embodiment mode, although the NAND and NOR have been describedwith reference to FIGS. 4A to 4D and FIGS. 5A to 5D, the presentinvention is applicable to other circuits. However, it is preferablethat the present invention is applied to a circuit that uses at leasttwo signals.

Embodiment Mode 5

This embodiment mode according to the present invention will bedescribed with reference to FIGS. 8A to 8C.

FIG. 8A shows appearance of a display device. The display device has apixel portion 102 in which (x×y) pixels 101 are arranged in a matrix ona substrate 107. A signal line driver circuit 103, a first scanning linedriver circuit 104 and a second scanning line driver circuit 105 arearranged on the periphery of the pixel portion 102. A signal isexternally supplied to the signal line driver circuit 103, the firstscanning line driver circuit 104, and the second scanning line drivercircuit 105 through a FPC 106. In addition, the signal line drivercircuit 103, the first scarring line driver circuit 104 and the secondscanning line driver circuit 105 may be provided outside the substrate107 in which the pixel portion 102 is formed. In FIG. 8A, one signalline driver circuit and two scanning driver circuits are provided, butthe numbers of signal line driver circuit and scanning line drivercircuit are not limited thereto. The numbers of them can be setarbitrarily corresponding to a structure of the pixel 101. Note that adisplay device in the present invention includes a panel in which apixel portion and a driver circuit are sealed between a substrate and acover material, a module in which an IC and the like are mounted on thepanel, and a display.

FIG. 8B shows an example of a structure of the signal line drivercircuit 103. The signal line driver circuit 103 has a shift register111, a first latch circuit 112, and a second latch circuit 113. FIG. 8Cshows an example of a structure of the first scanning line drivercircuit 104. The first scanning line driver circuit 104 has a shiftregister 114 and a buffer 115. Any one of the structures shown in FIGS.1A to 3D, 6A to 7B is freely applied to the shift register 111 or theshift register 114. Any one of the structures shown in FIGS. 4A to 5D oranother circuit according to the present invention is freely applied tothe first latch circuit 112, the second latch circuit 113 or the buffer115.

This embodiment mode can be freely combined with Embodiment Modes 1 to4.

Embodiment Mode 6

The following are examples of electronic appliances to which the presentinvention is applied: video cameras, digital cameras, goggle typedisplays (head mounted display), navigation systems, audio playbackunits (car audios, audio components, etc.), notebook type personalcomputers, game machines, portable information terminals (mobilecomputers, mobile telephones, mobile type game machines, electronicbooks, etc.), image playback units equipped with a recording medium(specifically, devices equipped with displays each of which is capableof playing a recording medium such as a digital versatile disk (DVD) anddisplaying the image thereof), and the like.

FIG. 9A shows a light emitting device, which includes a casing 2001, asupport base 2002, a display portion 2003, a speaker portion 2004, avideo input terminal 2005 and the like. The present invention can beapplied to a driver circuit of the display portion 2003. The lightemitting device shown in FIG. 9A can be completed according to thepresent invention. The light emitting device have a thinner displayportion than a liquid crystal display device, since the light emittingdevice is a self-luminous and does not need a backlight. Note that alldisplay devices for display information, for example, personalcomputers, devices for receiving TV broadcasting, and devices fordisplaying advertising, are also included in the light emitting device.

FIG. 9B shows a digital still camera, which includes a main body 2101, adisplay portion 2102, an image-receiving portion 2103, operation keys2104, an external connection port 2105, a shutter 2106 and the like. Thepresent invention can be applied to a driver circuit of the displayportion 2102. The digital still camera shown in FIG. 9B is completedaccording to the present invention.

FIG. 9C shows a notebook type personal computer, which includes a mainbody 2201, a casing 2202, a display portion 2203, a keyboard 2204,external connection ports 2205, a pointing mouse 2206, and the like. Thepresent invention can be applied to a driver circuit of the displayportion 2203. The notebook type personal computer shown in FIG. 9C iscompleted according to the present invention.

FIG. 9D shows a mobile computer, which includes a main body 2301, adisplay portion 2302, switches 2303, operation keys 2304, an infraredport 2305, and the like. The present invention can be applied to adriver circuit of the display portion 2302. The mobile computer shown inFIG. 9D is completed according to the present invention.

FIG. 9E shows a portable image playback unit provided with a recordingmedium (specifically, a DVD player), which includes a main body 2401, acasing 2402, a display portion A 2403, a display portion B 2404, arecording medium (such as a DVD) read-in portion 2405, operation keys2406, a speaker portion 2407, and the like. The display portion A 2403mainly displays image information, and the display portion B 2404 mainlydisplays character information. The present invention can be applied todriver circuits of the display portions A 2403 and B 2402. Note thatimage playback units provided with a recording medium include gamemachines for domestic use or the like. The image playback unit shown inFIG. 9E are completed according to the present invention.

FIG. 9F shows a goggle type display (head mounted display), whichincludes a main body 2501, a display portion 2502, an arm portion 2503,and the like. The present invention can be applied to a driver circuitof the display portion 2502. The goggle type display shown in FIG. 9F iscompleted according to the present invention.

FIG. 9G shows a video camera, which includes a main body 2601, a displayportion 2602, a casing 2603, external connection ports 2604, aremote-controlled receiving portion 2605, an image receiving portion2606, a battery 2607, an audio input portion 2608, operation keys 2609,an eye piece 2610, and the like. A pixel portion provided with a lightemitting element formed according to the present invention may beapplied to the display portion 2602. The video camera shown in FIG. 9Gis completed according to the present invention.

FIG. 9H shows a mobile telephone, which includes a main body 2701, acasing 2702, a display portion 2703, an audio input portion 2704, anaudio output portion 2705, operation keys 2706, external connectionports 2707, an antenna 2708, and the like. The present invention can beapplied to a driver circuit of the display portion 2703. Note that bydisplaying white characters on a black background in the display portion2703, the power consumption of the mobile telephone can be reduced. Themobile phone shown in FIG. 9H is completed according to the presentinvention.

In addition, miniaturization of casings for electronic appliances,reduction of an area occupied by a driver circuit in an internalcircuit, reduction of manufacturing costs, reduction of powerconsumption, and a high frequency operation are realized according tothe present invention. The present invention can give synergisticeffects to all the above electronic appliances and further, greatersynergistic effects to mobile terminals in particular.

As described above, the present invention can be widely applied to andused in electronic appliances in various fields. Further, the electronicappliances of this embodiment mode may employ any one of the pixelstructures of Embodiment Modes 1 to 5.

Embodiment Mode 7

A seventh structure of the present invention will be described withreference to FIGS. 12A and 12B. FIG. 12A is a circuit diagram showingstructural elements of a shift register arranged in the n-th stage. Eachstage is formed by a clocked inverter 10 including TFTs 11 and 13connected in series, a compensation circuit 19 a including TFTs 14 a and15 a, a compensation circuit 19 b including TFTs 14 b and 15 b, aholding clocked inverter 17 including TFTs 24 and 181 connected inseries, a compensation circuit including TFT 182 and an analog switch184, and a compensation circuit including TFT 183 and an analog switch185. The shift register is formed by cascade-connecting the respectivestages, in which these circuits are arranged, with signals from CK andCKB being alternately inputted at the respective stages. This structurein FIG. 12A is different from the structure in FIG. 3A in that CKB isinput to the source of TFT 14 a instead of inputting two-stage-beforesignal, TFT 181 is arranged in the holding clocked inverter 17 insteadof arranging TFTs 22 and 23, the compensation circuit comprising the TFT182 and the analog switch 184 is connected to a gate of the TFT 181, andthe compensation circuit comprising the TFT 183 and the analog switch185 is connected to a gate of the TFT 24.

Operations during periods T1 to T3 will be described using a timingchart shown in FIG. 12B. In the period T1, VSS is output from theclocked inverter 10.

Next, the operation during the period T2 is described here. VDD is inputto a gate of the TFT 181 to turn off in the clocked inverter 17. The TFT24 is on-state. Accordingly, VSS is output as OUT. In addition, in theclocked inverter 10, the TFT 11 is on-state or off-state. Even if theTFT 11 is on-state, VSS is output stably as OUT during the period 12since the TFT 24 has a high current performance.

In the above structure, it is not necessary to use a two-stage-beforesignal as the structures shown in FIGS. 3A and 3C. Therefore, the numberof leading out wirings can be reduced. This structure can be combinedwith any one of the above structures.

In the present invention having the first or second structure, a TFT isturned on at a desired timing by using a two-stage-before signal.

In the present invention having the three or four structure, a timing atwhich a TFT of a compensation circuit is turned on is delayed and thus,a timing at which a leak current flows is delayed by connecting multipleinverters to an input terminal of the compensation circuit, even if athreshold voltage of the TFT of the compensation circuit is equal to orless than a desired value. On the other hand, the threshold voltage ofthe TFT of the compensation circuit is equal to or more than a desiredvalue, generation of a leak current can be suppressed.

In the present invention having the fifth or sixth structure, a currentperformance of a clocked inverter is set large to accurately hold asignal. Further, it is possible to supply a signal with a stablewaveform, not being blunted, when the signal rises or falls.

Further, in the present invention, it is possible to change adouble-gate TFT (two TFTs connected in series) into a single-gate TFT.As a result, in the present invention, it is not required to increasethe gate widths of the TFTs and it is possible to reduce the sizes ofthe TFTs, which makes it possible to realize a high integration.Further, a burden on an element, whose gate (gate capacitance) functionsas a load, is reduced and the overall load is also reduced, so that ahigh frequency operation becomes possible. It is also possible toenhance the current performance of the TFT to be used. An accurateoperation is performed with a low voltage, even when a signal with 3 Vamplitude is used directly, since the structures according to thepresent invention are not influenced by fluctuation in a thresholdvoltage of the TFT.

1. A semiconductor device comprising: a first transistor; a secondtransistor; a third transistor; a fourth transistor; a fifth transistor;and a sixth transistor, wherein: a gate of the first transistor iselectrically connected to one of a source and a drain of the thirdtransistor and one of a source and a drain of the fourth transistor, oneof a source and a drain of the first transistor and the other of thesource and the drain of the fourth transistor are electrically connectedto a first power source, the other of the source and the drain of thefirst transistor is electrically connected to one of a source and adrain of the second transistor, a gate of the second transistor iselectrically connected to one of a source and a drain of the fifthtransistor and one of a source and a drain of the sixth transistor, theother of the source and the drain of the second transistor and the otherof the source and the drain of the sixth transistor are electricallyconnected to a second power source, a gate of the third transistor iselectrically connected to a gate of the fourth transistor, and a gate ofthe fifth transistor is electrically connected to a gate of the sixthtransistor.
 2. A semiconductor device according to claim 1, wherein afirst signal is inputted to the other of the source and the drain of thethird transistor, and wherein a second signal is inputted to the otherof the source and the drain of the fifth transistor.
 3. A semiconductordevice according to claim 1, wherein a first signal is inputted to theother of the source and the drain of the third transistor, wherein asecond signal is inputted to the other of the source and the drain ofthe fifth transistor, and wherein the first signal is different from thesecond signal.
 4. A semiconductor device according to claim 1, whereineach of the first transistor, the second transistor, the thirdtransistor, the fourth transistor, the fifth transistor, and the sixthtransistor is a thin film transistor.
 5. A semiconductor deviceaccording to claim 1, further including a plurality of pixels.
 6. Asemiconductor device according to claim 1, further including a driver.7. A semiconductor device comprising: a first transistor; a secondtransistor; a third transistor; a fourth transistor; and a fifthtransistor, wherein: a gate of the first transistor is electricallyconnected to one of a source and a drain of the fourth transistor andone of a source and a drain of the fifth transistor, one of a source anda drain of the first transistor and the other of the source and thedrain of the fifth transistor are electrically connected to a firstpower source, the other of the source and the drain of the firsttransistor is electrically connected to one of a source and a drain ofthe second transistor, the other of the source and the drain of thesecond transistor is electrically connected to one of a source and adrain of the third transistor, the other of the source and the drain ofthe third transistor is electrically connected to a second power source,and a gate of the fourth transistor is electrically connected to a gateof the fifth transistor.
 8. A semiconductor device according to claim 7,wherein a first signal is inputted to the other of the source and thedrain of the fourth transistor, and wherein a second signal is inputtedto a gate of the second transistor.
 9. A semiconductor device accordingto claim 7, wherein a first signal is inputted to the other of thesource and the drain of the fourth transistor, wherein a second signalis inputted to a gate of the second transistor, and wherein the firstsignal is different from the second signal.
 10. A semiconductor deviceaccording to claim 7, wherein each of the first transistor, the secondtransistor, the third transistor, the fourth transistor, and the fifthtransistor is a thin film transistor.
 11. A semiconductor deviceaccording to claim 7, further including a plurality of pixels.
 12. Asemiconductor device according to claim 7, further including a driver.13. A semiconductor device comprising: a first transistor; a secondtransistor; a third transistor; a fourth transistor; a fifth transistor;a sixth transistor; and a seventh transistor, wherein: a gate of thefirst transistor is electrically connected to one of a source and adrain of the fourth transistor and one of a source and a drain of thefifth transistor, one of a source and a drain of the first transistorand the other of the source and the drain of the fifth transistor areelectrically connected to a first power source, the other of the sourceand the drain of the first transistor is electrically connected to oneof a source and a drain of the second transistor, the other of thesource and the drain of the second transistor is electrically connectedto one of a source and a drain of the third transistor, a gate of thethird transistor is electrically connected to one of a source and adrain of the sixth transistor and one of a source and a drain of theseventh transistor, the other of the source and the drain of the thirdtransistor and the other of the source and the drain of the seventhtransistor are electrically connected to a second power source, a gateof the fourth transistor is electrically connected to a gate of thefifth transistor, and a gate of the sixth transistor is electricallyconnected to a gate of the seventh transistor.
 14. A semiconductordevice according to claim 13, wherein a first signal is inputted to theother of the source and the drain of the fourth transistor, and whereina second signal is inputted to the other of the source and the drain ofthe sixth transistor.
 15. A semiconductor device according to claim 13,wherein a first signal is inputted to the other of the source and thedrain of the fourth transistor, wherein a second signal is inputted tothe other of the source and the drain of the sixth transistor, andwherein the first signal is different from the second signal.
 16. Asemiconductor device according to claim 13, wherein each of the firsttransistor, the second transistor, the third transistor, the fourthtransistor, the fifth transistor, the sixth transistor, and the seventhtransistor is a thin film transistor.
 17. A semiconductor deviceaccording to claim 13, further including a plurality of pixels.
 18. Asemiconductor device according to claim 13, further including a driver.